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mic26903 28v, 9a hyper light load ? synchronous dc/dc buck regulator superswitcher iig ? hyper speed control, hyper light load, superswitcher ii, and any capacitor are trademarks of micrel, inc. mlf and micro leadframe are registered trademarks of amkor technology, inc. micrel inc. 2180 fortune drive san jose, ca 95131 usa tel +1 ( 408 ) 944-0800 fax + 1 (408) 474-1000 http://www.micrel.com general description the micrel mic26903 is a constant-frequency, synchronous dc/dc buck regulat or featuring adaptive on- time control architecture. the mic26903 operates over a supply range of 4.5v to 28v. it has an internal linear regulator which provides a regulated 5v to power the internal control circuitry. mi c26903 operates at a constant 600khz switching frequency in continuous conduction mode and can be used to provide up to 9a of output current. the output voltage is adjustable down to 0.8v. micrels hyper light load ? architecture provides the same high-efficiency and ultra-fast transient response as the hyper speed control ? architecture under medium to heavy loads, but also maintains high efficiency under light load conditions by transitioning to variable frequency, discontinuous mode operation. the mic26903 offers a full suite of protection features to ensure protection of the ic dur ing fault conditions. these include undervoltage lockout to ensure proper operation under power-sag conditions, thermal shutdown, internal soft-start to reduce the inru sh current, foldback current limit and hiccup mode short-circuit protection. the mic26903 includes a power good (pg) output to allow simple sequencing. all support documentation can be found on micrels web site at: www.micrel.com . features ? hyper light load ? efficiency C up to 80% at 10ma ? hyper speed control ? architecture enables ? high delta v operation (v in = 28v and v out = 0.8v) ? small output capacitance ? input voltage range: 4.5v to 28v ? output current up to 9a ? up to 95% efficiency ? adjustable output voltage from 0.8v to 5.5v ? 1% fb accuracy ? any capacitor tm stable ? zero-to-high esr ? 600khz switching frequency ? power good (pg) output ? foldback current-limit and hiccup mode short-circuit protection ? safe start-up into pre-biased loads ? 5mm 6mm mlf ? package ? C40 c to +125 c junction temperature range applications ? distributed power systems ? telecom/networking infrastructure ? printers, scanners, graphic cards and video cards ___________________________________________________________________________________________________________ typical application efficiency (v in = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0369 output current (a) efficiency (%) 1 2 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v v in = 12v july 2011 m9999-071311-a downloaded from: http:///
micrel, inc. mic26903 july 2011 2 m9999-071311-a ordering information part number voltage switching frequency junction temperature range package lead finish mic26903yjl adjustable 600khz C40 c to +125 c 28-pin 5mm 6mm mlf ? pb-free pin configuration 28-pin 5mm x 6mm mlf ? (yjl) pin description pin number pin name pin function 1 pvdd 5v internal linear regulator (output): pv dd supply is the power mosfet gate drive supply voltage and created by internal ldo from v in . when v in < +5.5v, pvdd should be tied to pvin pins. a 2.2f ceramic capacitor from the pvdd pin to pgnd (pin 2) must be place next to the ic. 3 nc no connect. 4, 9, 10, 11, 12 sw switch node (output): internal connection for the high-side mosfet source and low-side mosfet drain. due to the high speed switching on this pin, the sw pin should be routed away from sensitive nodes. 2, 5, 6, 7, 8, 21 pgnd power ground. pgnd is the ground path for t he mic26903 buck converter power stage. the pgnd pins connect to the low-side n-channel internal mosfet gate drive supply ground, the sources of the mosfets, the negat ive terminals of input capacitors, and the negative terminals of output capacitors. the loop for the power ground should be as small as possible and separate from the signal ground (sgnd) loop. 13,14,15, 16,17,18,19 pvin high-side n-internal mosfet drain connection (input): the pv in operating voltage range is from 4.5v to 26v. input capacitors between the pvin pins and the power ground (pgnd) are required and keep the connection short. 20 bst boost (output): bootstrapped vo ltage to the high-side n-channel mosfet driver. a schottky diode is connected between the pvdd pin and th e bst pin. a boost capacitor of 0.1 f is connected between the bst pin and the sw pin. adding a small resistor at the bst pin can slow down the turn-on time of high-side n-channel mosfets. downloaded from: http:///
micrel, inc. mic26903 july 2011 3 m9999-071311-a pin description (continued) pin number pin name pin function 22 cs current sense (input): the cs pi n senses current by monitoring the voltage across the low-side mosfet during the off-time. the current sensing is necessary for short circuit protection and zero current cross comparator. in order to sense the current accurately, connect the low-side mosfet drain to sw using a kelvin connection. the cs pin is also the high-side mosfets output driver return. 23 sgnd signal ground. sgnd must be conn ected directly to the ground pla nes. do not route the sgnd pin to the pgnd pad on the top layer, see pcb layout guidelines for details. 24 fb feedback (input): input to the transconductance amplif ier of the control loop. the fb pin is regulated to 0.8v. a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 25 pg power good (output): open drain output. the pg pi n is externally tied with a resistor to vdd. a high output is asserted when v out > 92% of nominal. 26 en enable (input): a logic level control of the out put. the en pin is cmos-compatible. logic high = enable, logic low = shutdown. in the off state, suppl y current of the device is greatly reduced (typically 5a). the en pin should not be left open. 27 vin power supply voltage (input): requires bypass capacitor to sgnd. 28 vdd 5v internal linear regulator (output ): vdd supply is the supply bus for the ic control circuit. vdd is created by internal ldo from v in . when v in < + 5.5v, vdd should be tied to pvin pins. a 1.0f ceramic capacitor from the vdd pin to sgnd pins must be place next to the ic. downloaded from: http:///
micrel, inc. mic26903 july 2011 4 m9999-071311-a absolute maximum ratings (1,2) pv in to pgnd................................................ ? 0.3v to +29v v in to pgnd .................................................... ? 0.3v to pv in pv dd , v dd to pgnd ......................................... ? 0.3v to +6v v sw , v cs to pgnd .............................. ? 0.3v to (pv in +0.3v) v bst to v sw ........................................................ ? 0.3v to 6v v bst to pgnd .................................................. ? 0.3v to 35v v fb , v pg to pgnd............................... ? 0.3v to (v dd + 0.3v) v en to pgnd ........................................ ? 0.3v to (v in +0.3v) pgnd to sgnd ........................................... ? 0.3v to +0.3v junction temperature .............................................. +150c storage temperature (t s )......................... ? 65 c to +150 c lead temperature (solde ring, 10sec ) ........................ 260c operating ratings (3) supply voltage (pv in , v in )................................. 4.5v to 28v pvdd, vdd supply voltage (pv dd , v dd )......... 4.5v to 5.5v enable input (v en ) .................................................. 0v to v in junction temperature (t j ) ........................ ? 40 c to +125 c maximum power dissi pation......................................note 4 package thermal resistance (4) 5mm x 6mm mlf ? -24l ( ja ) .............................28 c/w electrical characteristics (5) pv in = v in = v en = 12v, v bst C v sw = 5v; t a = 25c, unless noted. bold values indicate ? 40c t j +125c. parameter condition min. typ. max. units power supply input input voltage range (v in , pv in ) 4.5 28 v quiescent supply current v fb = 1.5v (non-switching) 450 750 a shutdown supply current v en = 0v 5 10 a v dd supply voltage v dd output voltage v in = 7v to 28v, i dd = 25ma 4.8 5 5.4 v v dd uvlo threshold v dd rising 3.7 4.2 4.5 v v dd uvlo hysteresis 400 mv dropout voltage (v in C v dd ) i dd = 25ma 380 600 mv dc/dc controller output-voltage adjust range (v out ) 0.8 5.5 v reference 0 c t j 85 c (1.0%) 0.792 0.8 0.808 ? 40 c t j 125 c (1.5%) 0.788 0.8 0.812 v load regulation i out = 3a to 9a (continuous mode) 0.25 % line regulation v in = 4.5v to 28v 0.25 % fb bias current v fb = 0.8v 50 500 na enable control en logic level high 1.8 v en logic level low 0.6 v en bias current v en = 12v 6 30 a oscillator switching frequency (6) 450 600 750 khz maximum duty cycle (7) v fb = 0v 82 % minimum duty cycle v fb = 1.0v 0 % minimum off-time 300 ns downloaded from: http:///
micrel, inc. mic26903 july 2011 5 m9999-071311-a electrical characteristics (5) (continued) pv in = v in = v en = 12v, v bst C v sw = 5v; t a = 25c, unless noted. bold values indicate ? 40c t j +125c. parameter condition min. typ. max. units soft-start soft-start time 5 ms short-circuit protection current-limit threshold v fb = 0.8v, t j = 25 c 12.5 15 20 a current-limit threshold v fb = 0.8v, t j = 125 c 11.25 15 20 a short-circuit current v fb = 0v 4 a internal fets top-mosfet r ds (on) i sw = 3a 27 m ? bottom-mosfet r ds (on) i sw = 3a 10.5 m ? sw leakage current v en = 0v 60 a v in leakage current v en = 0v 25 a power good (pg) pg threshold voltage sweep v fb from low to high 85 92 95 %v out pg hysteresis sweep v fb from high to low 5.5 %v out pg delay time sweep v fb from low to high 100 s pg low voltage sweep v fb < 0.9 v nom , i pg = 1ma 70 200 mv thermal protection over-temperature shutdown t j rising 160 c over-temperature shutdown hysteresis 15 c notes: 1. exceeding the absolute maximum rating may damage the device. 2. devices are esd sensitive. handling pr ecautions recommended. human body model, 1.5k ? in series with 100pf. 3. the device is not guaranteed to function outside operating range. 4. pd (max) = (t j(max) C t a )/ ja , where ja depends upon the printed circuit layout. a 5 square inch 4 layer, 0.62, fr-4 pcb with 2oz finish copper weight per layer is used for the ja . 5. specification for packaged product only. 6. measured in test mode. 7. the maximum duty-cycle is limited by the fixed mandatory off-time t off of typically 300ns. downloaded from: http:///
micrel, inc. mic26903 july 2011 6 m9999-071311-a typical characteristics v in operating supply current vs. input voltage 0.0 0.2 0.4 0.6 0.8 1.0 4 1 01 62 22 8 v in shutdown current vs. input voltage 0 15 30 45 60 4 1 01 62 22 8 v dd output voltage vs. input voltage 0 2 4 6 8 10 4 1 01 62 22 input voltage (v) v dd voltage (v) input voltage (v) shutdown current (a) v en = 0v r en = open input voltage (v) supply current (ma) v out = 1.8v i out = 0a sw itching 8 v fb = 0.9v i dd = 10ma feedback voltage vs. input voltage 0.792 0.796 0.800 0.804 0.808 4 1 01 62 22 8 total regulation vs. input voltage -0.2% -0.1% 0.0% 0.1% 0.2% 4 1 01 62 22 8 current limit vs. input voltage 0 4 8 12 16 20 4 1 01 62 22 input voltage (v) current limit (a) input voltage (v) feedback voltage (v) input voltage (v) total regulation (%) v out = 1.8v i out = 2a to 9a v out = 1.8v i out = 2a 8 v out = 1.8v switching frequency vs. input voltage 500 550 600 650 700 4 1 01 62 22 8 enable input current vs. input voltage 0 4 8 12 16 4 1 01 62 22 8 pg/v ref ratio vs. input voltage 80% 85% 90% 95% 100% 4.0 10.0 16.0 22.0 28.0 input voltage (v) v pg threshold/v ref (%) v ref = 0.7v input voltage (v) en input current (a) v en = v in input voltage (v) frequency (khz) v out = 1.8v i out = 2a downloaded from: http:///
micrel, inc. mic26903 july 2011 7 m9999-071311-a typical characteristics (continued) v in operating supply current vs. temperature 0.0 0.2 0.4 0.6 0.8 1.0 -50 -25 0 25 50 75 100 125 temperature (c) supply current (ma) v in = 12v v out = 1.8v i out = 0a sw itching v in shutdown current vs. temperature 0 2 4 6 8 10 -50 -25 0 25 50 75 100 125 temperature (c) shutdown current (ua) v in = 12v i out = 0a v en = 0v v dd uvlo threshold vs. temperature 0 1 2 3 4 5 -50 -25 0 25 50 75 100 125 temperature (c) v dd threshold (v) rising falling hyst feedback voltage vs. temperature 0.788 0.792 0.796 0.800 0.804 0.808 -50 -25 0 25 50 75 100 125 temperature (c) feeback voltage (v) v in = 12v v out = 1.8v i out = 2a load regulation vs. temperature -0.2% -0.1% 0.0% 0.1% 0.2% -50 -25 0 25 50 75 100 125 temperature (c) load regulation (%) v in = 12v v out = 1.8v i out =2a to 9a line regulation vs. temperature -0.2% -0.1% 0.0% 0.1% 0.2% -50 -25 0 25 50 75 100 125 temperature (c) line regulation (%) v in = 4.5v to 28v v out = 1.8v i out = 2a switching frequency vs. temperature 500 550 600 650 700 -50 -25 0 25 50 75 100 125 temperature (c) frequency (khz) v in = 12v v out = 1.8v i out = 2a v dd vs. temperature 2 3 4 5 6 -50 -25 0 25 50 75 100 125 temperature (c) vdd (v) v in = 12v v out = 1.8v i out =0a current limit vs. temperature 0 4 8 12 16 20 -50 -25 0 25 50 75 100 125 temperature (c) current limit (a) v in = 12v v out = 1.8v downloaded from: http:///
micrel, inc. mic26903 july 2011 8 m9999-071311-a typical characteristics (continued) efficiency vs. output current 50 60 70 80 90 100 01 . 534 . 567 . 59 output current (a) efficiency (%) 12v in 24v in v out = 1.8v feedback voltage vs. output current 0.792 0.796 0.800 0.804 0.808 0 1.5 3 4.5 6 7.5 9 output current (a) feedback voltage (v) v in = 12v v out = 1.8v output voltage vs. output current 1.782 1.787 1.791 1.796 1.800 1.805 1.810 1.814 1.819 01.534.567.59 output current (a) output voltage (v) v in = 12v v out = 1.8v line regulation vs. output current -1.0% -0.5% 0.0% 0.5% 1.0% 0235689 output current (a) line regulation (%) v in = 4.5v to 28v v out = 1.8v switching frequency vs. output current 500 550 600 650 700 34 . 567 . 5 output current (a) frequency (khz) 9 v in = 12v v out = 1.8v output voltage (v in = 5v) vs. output current 3.0 3.4 3.8 4.2 4.6 5.0 024681 01 output current (a) output voltage (v) 2 t a 25oc 85oc 125oc v in = 5v v fb < 0.8v efficiency (v in = 5v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 03691 output current (a) efficiency (%) 2 ic power dissipation (vin = 5v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 01 . 534 . 567 . 59 output current (a) power dissipation (w) v in = 5v v out = 0.8,v, 1.0v, 1.2v, 1.5v, 1.8v,2.5v, 3.3v 3.3v 0.8v die temperature* (v in = 5v) vs. output current 0 20 40 60 80 100 023568 output current (a) die temperature (c) 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v 9 v in = 5v v out = 1.8v v in = 5v downloaded from: http:///
micrel, inc. mic26903 july 2011 9 m9999-071311-a typical characteristics (continued) efficiency (v in = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 03691 output current (a) efficiency (%) 2 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v v in = 12v ic power dissipation (vin = 12v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 01 . 534 . 567 . 59 output current (a) power dissipation (w) v in = 12v v out = 0.8,v, 1.0v, 1.2v, 1.5v, 1.8v,2.5v, 3.3v, 5.0v 5.0v 0.8v die temperature* (v in = 12v) vs. output current 0 20 40 60 80 100 0235689 output current (a) die temperature (c) v in = 12v v out = 1.8v efficiency (v in = 24v) vs. output current 50 55 60 65 70 75 80 85 90 95 03691 output current (a) efficiency (%) 2 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v v in = 24v ic power dissipation (vin = 24v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 01 . 534 . 567 . 59 output current (a) power dissipation (w) v in = 24v v out = 0.8,v, 1.0v, 1.2v, 1.5v, 1.8v,2.5v, 3.3v, 5.0v 5.0v 0.8v die temperature* (v in = 24v) vs. output current 0 20 40 60 80 100 0235689 output current (a) die temperature (c) v in = 24v v out = 1.8v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50-25 0 25 50 75100125 ambient temperature (c) output current (a) 1.5v 0.8v v in = 5v v out = 0.8, 1.2, 1.5v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) 3.3v 1.8v v in = 5v v out = 1.8, 2.5, 3.3v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) v in = 12v v out = 0.8, 1.2, 1.8v 1.8v 0.8v downloaded from: http:///
micrel, inc. mic26903 july 2011 10 m9999-071311-a typical characteristics (continued) thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50-25 0 25 50 75100125 ambient temperature (c) output current (a) v in = 12v v out = 2.5, 3.3, 5v 5v 2.5v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) v in = 24v v out = 0.8, 1.2, 2.5v 2.5v 0.8v die temperature* : the temperature measurement was taken at the hottest point on the mic26903 case mounted on a 5 square inch 4 layer, 0.62, fr-4 pcb with 2oz finish copper weight per layer, see therma l measurement section. actual results will depend upon the size of the pcb, ambient temperature and proximity to other heat emitting components. downloaded from: http:///
micrel, inc. mic26903 july 2011 11 m9999-071311-a functional characteristics downloaded from: http:///
micrel, inc. mic26903 july 2011 12 m9999-071311-a functional characteristics (continued) downloaded from: http:///
micrel, inc. mic26903 july 2011 13 m9999-071311-a functional characteristics (continued) downloaded from: http:///
micrel, inc. mic26903 july 2011 14 m9999-071311-a functional diagram figure 1. mic26903 block diagram downloaded from: http:///
micrel, inc. mic26903 july 2011 15 m9999-071311-a functional description the mic26903 is an adaptive on-time synchronous step-down dc/dc regulator with an internal 5v linear regulator and a power good (pg) output. it is designed to operate over a wide input voltage range from 4.5v to 28v and provides a regulated output voltage at up to 7a of output current. an adaptive on-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. over-current protection is implemented without the use of an external sense resistor. the device includes an internal soft-start function which reduces the power supply input surge current at start-up by cont rolling the output voltage rise time. theory of operation the mic26903 is able to operate in either continuous mode or discontinuous mode. the operating mode is determined by the output of the zero cross comparator (zc) as shown in figure 1. continuous mode in continuous mode, the output voltage is sensed by the mic26903 feedback pin fb via the voltage divider r1 and r2, and compared to a 0.8v reference voltage v ref at the error comparator through a low gain transconductance (g m ) amplifier. if the feedback voltage decreases and the output of the g m amplifier is below 0.8v, then the error comparator will trigger the control logic and generate an on-time period. the on-time period length is predetermined by the fixed t on estimation circuitry: 600khz v v = t in out ed) on(estimat eq. 1 where v out is the output voltage and v in is the power stage input voltage. at the end of the on-time period, the internal high-side driver turns off the high-side mosfet and the low-side driver turns on the low-side mosfet. the off-time period length depends upon the feedback voltage in most cases. when the feedback voltage decreases and the output of the g m amplifier is below 0.8v, the on-time period is triggered and the off-time period ends. if the off-time period determined by the feedback voltage is less than the minimum off-time t off(min) , which is about 300ns, the mic26903 control logic will apply the t off(min) instead. t off(min) is required to maintain enough energy in the boost capacitor (c bst ) to drive the high-side mosfet. the maximum duty cycle is obtained from the 300ns t off(min) : s s off(min) s max t 300ns 1 t t t d - - = = eq. 2 where t s = 1/600khz = 1.66 s. it is not recommended to use mic26903 with a off-time close to t off(min) during steady-state operation. also, as v out increases, the internal ri pple injection will increase and reduce the line regulation performance. therefore, the maximum output voltage of the mic26903 should be limited to 5.5v and the maximum external ripple injection should be limited to 200mv. please refer to setting output voltage subsection in application information for more details. the actual on-time and re sulting switching frequency will vary with the part-to-part variation in the rise and fall times of the internal mosfets, the output load current, and variations in the v dd voltage. also, the minimum t on results in a lower switching frequency in high v in to v out applications, such as 24v to 1.0v. the minimum t on measured on the mic26903 evaluation board is about 100ns. during load transients, the switching frequency is changed due to the varying off-time. to illustrate the control lo op operation, we will analyze both the steady-state and load transient scenarios. figure 2 shows the mic26903 control loop timing during steady-state operation. during steady-state, the g m amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the on-time period. the on- time is predetermined by the t on estimator. the termination of the off-time is controlled by the feedback voltage. at the valley of the feedback voltage ripple, which occurs when v fb falls below v ref , the off period ends and the next on-time period is triggered through the control logic circuitry. downloaded from: http:///
micrel, inc. mic26903 july 2011 16 m9999-071311-a figure 2. mic26903 control loop timing figure 3 shows the operation of the mic26903 during a load transient. the output voltage drops due to the sudden load increase, which causes the v fb to be less than v ref . this will cause the error comparator to trigger an on-time period. at the end of the on-time period, a minimum off-time t off(min) is generated to charge c bst since the feedback voltage is still below v ref . then, the next on-time period is triggered due to the low feedback voltage. therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. with the va rying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in mic26903 converter. figure 3. mic26903 load transient response unlike true current-mode cont rol, the mic26903 uses the output voltage ripple to trigger an on-time period. the output voltage ripple is proportional to the inductor current ripple if the esr of the output capacitor is large enough. the mic26903 control loop has the advantage of eliminating the need for slope compensation. in order to meet the stability requirements, the mic26903 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the g m amplifier and the error comparator. the recommended feedback voltage ripple is 20mv~100mv. if a low-esr output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the g m amplifier and the error comparator. also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the esr of the output capacitor is very low. in these cases, ripple in jection is required to ensure proper operation. please refer to ripple injection subsection in application information for more details about the ripple injection technique. discontinuous mode in continuous mode, the inductor current is always greater than zero; however, at light loads the mic26903 is able to force the inductor current to operate in discontinuous mode. discontinuous mode is where the inductor current falls to zero, as indicated by trace (i l ) shown in figure 4. during this period, the efficiency is optimized by shutting down all the non-essential circuits and minimizing the supply current. the mic26903 wakes up and turns on the high-side mosfet when the feedback voltage v fb drops below 0.8v. the mic26903 has a zero crossing comparator that monitors the inductor current by sensing the voltage drop across the low-side mosfet during its on-time. if the v fb > 0.8v and the inductor current goes slightly negative, then the mic26903 automatically powers down most of the ic circuitry and goes into a low-power mode. once the mic26903 goes into discontinuous mode, both lsd and hsd are low, which turns off the high-side and low-side mosfets. the load current is supplied by the output capacitors and v out drops. if the drop of v out causes v fb to go below v ref , then all the circuits will wake up into normal continuous mode. first, the bias currents of most circuits reduced during the discontinuous mode are restored, then a t on pulse is triggered before the drivers are turned on to avoid any possible glitches. finally, the high-side driver is turned on. figure 4 shows the control loop timing in discontinuous mode. downloaded from: http:///
micrel, inc. mic26903 july 2011 17 m9999-071311-a figure 4. mic26903 control loop timing (discontinuous mode) during discontinuous mode, the zero crossing comparator and the current limit comparator are turned off. the bias current of most circuits are reduced. as a result, the total power supply current during discontinuous mode is only about 450 a, allowing the mic26903 to achieve high efficiency in light load applications. v dd regulator the mic26903 provides a 5v regulated output for input voltage v in ranging from 5.5v to 28v. when v in < 5.5v, v dd should be tied to pvin pins to bypass the internal linear regulator. soft-start soft-start reduces the power supply input surge current at startup by cont rolling the output voltage rise time. the input surge appears while the output capacitor is charged up. a slower output ri se time will draw a lower input surge current. the mic26903 implements an internal digital soft-start by making the 0.8v reference voltage v ref ramp from 0 to 100% in about 5ms with 9. 7mv steps. therefore, the output voltage is controlled to increase slowly by a stair- case v fb ramp. once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. v dd must be powered up at the same time or after v in to make the soft-start function correctly. current limit the mic26903 uses the r ds(on) of the internal low-side power mosfet to sense over -current conditions. this method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. the low-side mosfet is used because it displays much lower parasitic oscillations during switching than the high-side mosfet. in each switching cycle of the mic26903 converter, the inductor current is sensed by monitoring the low-side mosfet in the off period. if the inductor current is greater than 15a, then the mic26903 turns off the high- side mosfet and a soft-start sequence is triggered. this mode of operation is called hiccup mode and its purpose is to protect the downstream load in case of a hard short. the load current-limit threshold has a fold back characteristic related to the feedback voltage as shown in figure 5. power-good (pg) the power good (pg) pin is an open drain output which indicates logic high when the output is nominally 92% of its steady state voltage. a pull- up resistor of more than 10k ? should be connected from pg to vdd. current limit threshold vs. feedback voltage 0 4 8 12 16 20 0.0 0.2 0.4 0.6 0.8 1.0 feedback voltage (v) current limit threshold (a) figure 5. mic26903 current limit foldback characteristic downloaded from: http:///
micrel, inc. mic26903 july 2011 18 m9999-071311-a mosfet gate drive the block diagram (figure 1) shows a bootstrap circuit, consisting of d1 (a schottky diode is recommended) and c bst . this circuit supplies energy to the high-side drive circuit. capacitor c bst is charged, while the low-side mosfet is on, and the voltage on the sw pin is approximately 0v. when the high-side mosfet driver is turned on, energy from c bst is used to turn the mosfet on. as the high-side mosfet turns on, the voltage on the sw pin increases to approximately v in . diode d1 is reverse biased and c bst floats high while continuing to keep the high-side mosfet on. the bias current of the high-side driver is less than 10ma so a 0.1 f to 1 f is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. bst = 10ma x 1.67 s/0.1 f = 167mv. when the low-side mosfet is turned back on, c bst is recharged through d1. a small resistor r g , which is in series with c bst , can be used to slow down the turn-on time of the high-side n-channel mosfet. the drive voltage is derived from the v dd supply voltage. the nominal low-side gate drive voltage is v dd and the nominal high-side gate drive voltage is approximately v dd C v diode , where v diode is the voltage drop across d1. an approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both mosfets. downloaded from: http:///
micrel, inc. mic26903 july 2011 19 m9999-071311-a application information inductor selection values for inductance, peak, and rms currents are required to select the output inductor. the input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. generally, higher inductance values are used with higher input voltages. larger peak-to-peak ripple currents will increase the power dissipation in the inductor and mosfets. larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. a good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is calculated by equation 3: out(max) sw in(max) out in(max) out i 20% f v ) v (v v l ? = eq. 3 where: f sw = switching frequency, 600khz 20% = ratio of ac ripple current to dc output current v in(max) = maximum power stage input voltage the peak-to-peak inductor current ripple is: l f v ) v (v v i sw in(max) out in(max) out l(pp) ? = eq. 4 the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. i l(pk) =i out(max) + 0.5 i l(pp) eq. 5 the rms inductor current is used to calculate the i 2 r losses in the inductor. 12 i i i 2 l(pp) 2 out(max) l(rms) + = eq. 6 maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. the high frequency operation of the mic26903 requires the use of ferrite materials for all but the most cost sensitive applications. lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. this is especially noticeable at low output power. the windi ng resistance decreases efficiency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper losses. at higher output loads, the core losses are usually insignificant and can be ignored. at lower output currents, the core losses can be a significant contributor. core loss information is usually available from the magnetics vendor. copper loss in the inductor is calculated by equation 7: p inductor(cu) = i l(rms) 2 r winding eq. 7 the resistance of the copper wire, r winding , increases with the temperature. the value of the winding resistance used should be at the operating temperature. p winding(ht) = r winding(20c) (1 + 0.0042 (t h C t 20c )) eq. 8 where: t h = temperature of wire under full load t 20c = ambient temperature r winding(20c) = room temperature winding resistance (usually specified by the manufacturer) output capacitor selection the type of the output capacito r is usually determined by its equivalent series resistance (esr). voltage and rms current capability are two ot her important factors for selecting the output capacitor. recommended capacitor types are tantalum, low-esr aluminum electrolytic, os- con and poscap. the output capacitors esr is usually the main cause of the output ripple. the output capacitor esr also affects the control loop from a stability point of view. downloaded from: http:///
micrel, inc. mic26903 july 2011 20 m9999-071311-a the maximum value of esr is calculated: l(pp) out(pp) c i v esr out eq. 9 where: v out(pp) = peak-to-peak output voltage ripple i l(pp) = peak-to-peak inductor current ripple the total output ripple is a combination of the esr and output capacitance. the total ripple is calculated in equation 10: ( ) 2 c l(pp) 2 sw out l(pp) out(pp) out esr i 8 f c i v + ? ?? ? ? ?? ? = eq. 10 where: d = duty cycle c out = output capacitance value f sw = switching frequency as described in the theory of operation subsection in the functional description section, the mic26903 requires at least 20mv peak-to-peak ripple at the fb pin to make the g m amplifier and the error comparator behave properly. also, the output voltage ripple should be in phase with the inductor current. therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor esr. if low-esr capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. please refer to the ripple injection subsection for more details. the voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or os-con. the output capacitor rms current is calculated below: 12 i i l(pp) (rms) c out = eq. 11 the power dissipated in the output capacitor is: out out out c 2 (rms) c ) diss(c esr i p = eq. 12 input capacitor selection the input capacitor for the power stage input v in should be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. a tantalum input capacitors voltage rating should be at least two times the maximum input voltage to maximize reliability. aluminum electrolytic, os-con, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. the input voltage ripple will primarily depend on the input capacitors esr. the peak input current is equal to the peak inductor current, so: v in = i l(pk) esr cin eq. 13 the input capacitor must be rated for the input current ripple. the rms value of input capacitor current is determined at the maximum output current. assuming the peak-to-peak inductor current ripple is low: d) (1 d i i out(max) cin(rms) ? eq. 14 the power dissipated in the input capacitor is: p diss(cin) = i cin(rms) 2 esr cin eq. 15 ripple injection the v fb ripple required for proper operation of the mic26903 g m amplifier and error comparator is 20mv to 100mv. however, the output voltage ripple is generally designed as 1% to 2% of the output voltage. for a low output voltage, such as a 1v, the output voltage ripple is only 10mv to 20mv, and the feedback voltage ripple is less than 20mv. if the feedback voltage ripple is so small that the g m amplifier and error comparator cant sense it, then the mic26903 will lose control and the output voltage is not regulated. in order to have some amount of v fb ripple, a ripple injection method is applied for low output voltage ripple applications. downloaded from: http:///
micrel, inc. mic26903 july 2011 21 m9999-071311-a the applications are divided into three situations according to the amount of the feedback voltage ripple: 1. enough ripple at the feedback voltage due to the large esr of the output capacitors. as shown in figure 6a, the converter is stable without any ripple injection. the feedback voltage ripple is: (pp) l c fb(pp) i esr r2 r1 r2 v out + = eq. 16 where: i l(pp) is the peak-to-peak value of the inductor current ripple. 2. inadequate ripple at the feedback voltage due to the small esr of the output capacitors. the output voltage ripple is fed into the fb pin through a feedforward capacitor c ff in this situation, as shown in figure 6b. the typical c ff value is between 1nf and 100nf. with the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: (pp) l fb(pp) i esr v eq. 17 3. virtually no ripple at the fb pin voltage due to the very low esr of the output capacitors. figure 6a. enough ripple at fb figure 6b. inadequate ripple at fb mic26903 figure 6c. invisible ripple at fb in this situation, the output voltage ripple is less than 20mv. therefore, additional ripple is injected into the fb pin from the switching node sw via a resistor r inj and a capacitor c inj , as shown in figure 6c. the injected ripple is: f 1 d)-(1d k v= v sw div in fb(pp) eq. 18 r1//r2 r r1//r2 k inj div + = eq. 19 where: v in = power stage input voltage d = duty cycle f sw = switching frequency = (r1//r2//r inj ) c ff in equations 21 and 22, it is assumed that the time constant associated with c ff must be much greater than the switching period: mic26903 1 << t = f 1 sw eq. 20 if the voltage divider resistors r1 and r2 are in the k ? range, a c ff of 1nf to 100nf can easily satisfy the large time constant requirements. also, a 100nf injection capacitor c inj is used in order to be considered as short for a wide range of the frequencies. mic26903 downloaded from: http:///
micrel, inc. mic26903 july 2011 22 m9999-071311-a the process of sizing the ripple injection resistor and capacitors is: step 1. select c ff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. typical choice of c ff is 1nf to 100nf if r1 and r2 are in k ? range. step 2. select r inj according to the expected feedback voltage ripple using equation 22, d) (1 d f v v k sw in fb(pp) div - = eq. 21 then the value of r inj is obtained as: 1) k 1 ( (r1//r2) r div inj ? = eq. 22 step 3. select c inj as 100nf, which could be considered as short for a wide range of the frequencies. setting output voltage the mic26903 requires two resistors to set the output voltage as shown in figure 7. the output voltage is determined by the equation: ) r2 r1 (1 v v fb out + = eq. 23 where: v fb = 0.8v. a typical value of r1 can be between 3k ? and 10k ? . if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small, it will decrease the efficiency of the power supply, especially at light loads. once r1 is selected, r2 can be calculated using: fb out fb v v r1 v r2 ? = eq. 24 figure 7. voltage-divider configuration in addition to the external ripple injection added at the fb pin, internal ripple injection is added at the inverting input of the comparator in side the mic26903, as shown in figure 8. the inverting input voltage v inj is clamped to 1.2v. as v out is increased, the swing of v inj will be clamped. the clamped v inj reduces the line regulation because it is reflected as a dc error on the fb terminal. therefore, the maximum output voltage of the mic26903 should be limited to 5.5v to avoid this problem. figure 8. internal ripple injection downloaded from: http:///
micrel, inc. mic26903 july 2011 23 m9999-071311-a thermal measurements measuring the ics case temperature is recommended to insure it is within its operating limits. although this might seem like a very elementary task, it is easy to get erroneous results. the most common mistake is to use the standard thermal couple that comes with a thermal meter. this thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. if a thermal couple wire is used, it must be constructed of 36 gauge wire or higher then (smaller wire size) to minimize the wire heat-sinking effect. in addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the ic. omega brand thermal couple (5sc-tt-k-36-36) is adequate for most applications. wherever possible, an infrared thermometer is recommended. the measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ics. however, a ir thermometer from optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. an optional stand makes it easy to hold the beam on the ic for long periods of time. downloaded from: http:///
micrel, inc. mic26903 july 2011 24 m9999-071311-a pcb layout guidelines warning!!! to minimize emi and output noise, follow these layout recommendations. pcb layout is critical to achieve reliable, stable and efficient performance. a ground plane is required to control emi and minimize the inductance in power, signal and return paths. the following guidelines should be followed to insure proper operation of the mic26903 regulator. ic ? a 2.2f ceramic capacitor, which is connected to the pvdd pin, must be located right at the ic. the pvdd pin is very noise sensitive and placement of the capacitor is very critical. use wide traces to connect to the pvdd and pgnd pins. ? a 1.0uf ceramic capacitor must be placed right between vdd and the signal ground sgnd. the sgnd must be connected directly to the ground planes. do not route the sgnd pin to the pgnd pad on the top layer. ? place the ic close to the point-of-load (pol). ? use fat traces to route the input and output power lines. ? signal and power grounds should be kept separate and connected at only one location. input capacitor ? place the input capacitor next. ? place the input capacitors on the same side of the board and as close to the ic as possible. ? keep both the pvin pin and pgnd connections short. ? place several vias to the ground plane close to the input capacitor ground terminal. ? use either x7r or x5r dielectric input capacitors. do not use y5v or z5u type capacitors. ? do not replace the ceramic input capacitor with any other type of capacitor. any type of capacitor can be placed in parallel with the input capacitor. ? if a tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. ? in hot-plug applications, a tantalum or electrolytic bypass capacitor must be used to limit the over- voltage spike seen on the input supply with power is suddenly applied. inductor ? keep the inductor connection to the switch node (sw) short. ? do not route any digital lines underneath or close to the inductor. ? keep the switch node (sw) away from the feedback (fb) pin. ? the cs pin should be connected directly to the sw pin to accurate sense the voltage across the low- side mosfet. ? to minimize noise, place a ground plane underneath the inductor. ? the inductor can be placed on the opposite side of the pcb with respect to the ic. it does not matter whether the ic or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. the input and output capacitors must be placed on the same side of the board as the ic. output capacitor ? use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. ? phase margin will change as the output capacitor value and esr changes. contact the factory if the output capacitor is different from what is shown in the bom. ? the feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. sensing a long high current load trace can degrade the dc load regulation. optional rc snubber ? place the rc snubber on either side of the board and as close to the sw pin as possible. downloaded from: http:///
micrel, inc. mic26903 july 2011 25 m9999-071311-a evaluation board schematic figure 9. schematic of mic26903 evaluation board (j11, r13, r15 are for testing purposes) downloaded from: http:///
micrel, inc. mic26903 july 2011 26 m9999-071311-a bill of materials item part number manufacturer description qty. c1 open 12105c475kaz2a avx (1) grm32er71h475ka88l murata (2) c2, c3 c3225x7r1h475k tdk (3) 4.7f ceramic capacitor, x7r, size 1210, 50v 2 c13, c15 open 12106d107mat2a avx (1) grm32er60j107me20l murata (2) c4, c5 c3225x5r0j107m tdk (3) 100f ceramic capacitor, x5r, size 1210, 6.3v 3 06035c104kat2a avx (1) grm188r71h104ka93d murata (2) c6, c7, c10 c1608x7r1h104k tdk (3) 0.1f ceramic capacitor, x7r, size 0603, 50v 3 0603zc105kat2a avx (1) grm188r71a105ka61d murata (2) c8 c1608x7r1a105k tdk (3) 1.0f ceramic capacitor, x7r, size 0603, 10v 1 0603zd225kat2a avx (1) grm188r61a225ke34d murata (2) c9 c1608x5r1a225k tdk (3) 2.2f ceramic capacitor, x5r, size 0603, 10v 1 06035c472kaz2a avx (1) grm188r71h472k murata (2) c12 c1608x7r1h472k tdk (3) 4.7nf ceramic capacitor, x7r, size 0603, 50v 1 c14 b41851f7227m epcos (4) 220f aluminum capacitor, 35v 1 c11, c16 open sd103aws mcc (5) sd103aws-7 diodes inc (6) d1 sd103aws vishay (7) 40v, 350ma, schottky diode, sod323 1 l1 hcf1305-2r2-r cooper bussmann (8) 2.2h inductor, 15a saturation current 1 r1 crcw06032r21fkea vishay dale (7) 2.21 ? resistor, size 0603, 1% 1 r2 crcw06032r00fkea vishay dale (7) 2.00 ? resistor, size 0603, 1% 1 r3 crcw060319k6fkea vishay dale (7) 19.6k ? resistor, size 0603, 1% 1 r4 crcw06032k49fkea vishay dale (7) 2.49k ? resistor, size 0603, 1% 1 r5 crcw060320k0fkea vishay dale (7) 20.0k ? resistor, size 0603, 1% 1 r6, r14, r17 crcw060310k0fkea vishay dale (7) 10.0k ? resistor, size 0603, 1% 3 r7 crcw06034k99fkea vishay dale (7) 4.99k ? resistor, size 0603, 1% 1 r8 crcw06032k87fkea vishay dale (7) 2.87k ? resistor, size 0603, 1% 1 r9 crcw06032k006fkea vishay dale (7) 2.00k ? resistor, size 0603, 1% 1 r10 crcw06031k18fkea vishay dale (7) 1.18k ? resistor, size 0603, 1% 1 r11 crcw0603806rfkea vishay dale (7) 806 ? resistor, size 0603, 1% 1 r12 crcw0603475rfkea vishay dale (7) 475 ? resistor, size 0603, 1% 1 downloaded from: http:///
micrel, inc. mic26903 july 2011 27 m9999-071311-a bill of materials (continued) item part number manufacturer description qty r13 crcw06030000fkea vishay dale (7) 0 ? resistor, size 0603, 5% 1 r15 crcw060349r9fkea vishay dale (7) 49.9 ? resistor, size 0603, 1% 1 r16, r18 crcw06031r21fkea vishay dale (7) 1.21 ? resistor, size 0603, 1% 2 r20 open u1 mic26903yjl micrel. inc. (9) 28v, 9a hyper light load ? synchronous dc/dc buck regulator 1 notes: 1. avx: www.avx.com . 2. murata: www.murata.com . 3. tdk: www.tdk.com . 4. epcos: www.epcos.com . 5. sanyo: www.sanyo.com . 6. diode inc.: www.diodes.com . 7. vishay: www.vishay.com . 8. cooper bussmann: www.cooperbussmann.com . 9. micrel, inc.: www.micrel.com . downloaded from: http:///
micrel, inc. mic26903 july 2011 28 m9999-071311-a pcb layout figure 10. mic26903 evaluation board top layer figure 11. mic26903 evaluation board mid-layer 1 (ground plane) downloaded from: http:///
micrel, inc. mic26903 july 2011 29 m9999-071311-a pcb layout (continued) figure 12. mic26903 evaluation board mid-layer 2 figure 13. mic26903 evaluation board bottom layer downloaded from: http:///
micrel, inc. mic26903 july 2011 30 m9999-071311-a recommended land and solder stencil pattern downloaded from: http:///
micrel, inc. mic26903 july 2011 31 m9999-071311-a package information 28-pin 5mm 6mm mlf ? (yjl) micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944-0800 fax +1 (408) 474-1000 web http://www.micrel.com micrel makes no representations or warranties with respect to t he accuracy or completeness of the information furnished in this data sheet. this information is not intended as a warranty and micrel does not assume responsibility for it s use. micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. no license, whether expre ss, implied, arising by estoppel or other wise, to any intellectual property rights is granted by this document. except as provided in micrels terms and conditions of sale for such products, mi crel assumes no liability whatsoever, and micrel disclaims any express or implied warranty relating to the sale and/or use of micrel products including l iability or warranties relating to fitness for a particular purpose, merchantability, or infringement of an y patent, copyright or other intellectual p roperty right. micrel products are not designed or authori zed for use as components in life support app liances, devices or systems where malfu nction of a product reasonably be expected to result in pers onal injury. life support devices or system s are devices or systems that (a) are in tended for surgical impla into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significan t injury to the user. a purchasers use or sale of micrel produc ts for use in life support app liances, devices or systems is a purchasers own risk and purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. can nt ? 2011 micrel, incorporated. downloaded from: http:///

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